Anal. Chem. 1980, 52, 1273-1278
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High Speed Pulse Amplifier/Discriminator and Counter for Photon Counting Richard A. Borders and John W. Birks" Department of Chemistry and Cooperative Institute for Research in Environmental Sciences (CIRES), University of Colorado, Boulder, Colorado 80309
John A. Borders Naval Ship Weapon Systems Engineering Station, Port Hueneme, California 93043
The design for a Pulse Ampllfier/Discrimlnator (PAD) and high speed counter is given. The PAD is completely self-contained, has complementary Emitter Coupled Logic (ECL) outputs which can drive a 5 0 4 load (coaxial cable) to a remote counter, and costs less than $150 to construct. The PAD is typically capable of detecting a single pulse with an amplitude of I100 y V , typically has a Pulse Pair Resolution (PPR) of 10 ns for 350-pV to I40-mV pulses, and can operate to >250 MHz with periodic Input pulses of I 8 mV. The front end of the counter circuit uses high speed ECL flipflops. The output data from the front end Is converted to TTL, and the remainder of the counting Is done using slower, cascaded l T L counters. The counter is capable of counting at frequencies >250 MHr, and the maximum count is 2= - 1. The counter is interfaced to a 8080-based microcomputer. The PAD and counter were tested using a matched pair of EM1 9658RA Photomultiplier Tubes (PMTs), and the relationship between current and photon counting was found to be linear within 5 % to >2 MHz.
Pulse (photon) counting has been widely compared with other methods of measuring the output signal of electron multipliers and found to be the method of choice for systems with low-to-moderate signal fluxes (1-1 I ) . T h e advantages of photon counting over analog measuring systems are (1) discrimination against noise which does not originate a t the photocathode, thereby increasing the system signal-to-noise ratio; (2) digital processing of inherently discrete spectral information; (3) elimination of the errors associated with data domain conversions; (4) sensitivity to very low light levels; ( 5 ) accurate long term integration; and (6) reduced sensitivity to l / fnoise, long term drift, voltage changes, and temperature changes. T h e disadvantage of photon counting is that pulse pileup occurs a t moderate-to-high signal fluxes causing the signal to become nonlinear (3-7, 12-15). Pulse pileup is caused by either the arrival of two or more photons a c h e photocathode within the pulse width of the first photon or by t h e arrival of photons at a rate which the amplifier/counting system cannot resolve. T h e measurement of high signal fluxes requires an analog measurement system or the use of dead time and/or count loss correction techniques (3,12-15) to extend t h e range of photon counting. T h e use of faster counting systems applies t h e advantages of photon counting to higher signal fluxes provided that the resolution of pulses is limited by the counting system, as is often the case. A block diagram of the pulse counting system is shown in Figure 1. T h e electron multiplier may be either the electron multiplier associated with a mass spectrometer or a PMT. In our system, the amplifier and discriminator are combined into a small (12 inch X 2 1 / 4 inch X 2 inch), self-contained module (the PAD). T h e output from the PAD is counted for a period of time which is controlled by a microcomputer. The microcomputer collects the data from the counters (and other ~~~~~
0003-2700/80/0352-1273$01 .OO/O
sensors), processes t h e data, and prints t h e result. T h e PAD and counter described in this article perform as well as, if not better than, commercially available systems a t a fraction of their cost.
EXPERIMENTAL Definition of Terms. Previous articles which describe amplifiers or amplifier/discriminators have not rigorously defined the terms which they have used to describe circuit behavior. This makes it extremely difficult to decide whether the circuits described are suitable for a particular application. To avoid this problem we define the following terms. False Alarm Rate. The false alarm rate is the number of false counts per second from the PAD when the input is terminated with its characteristic impedance (50 Q). The false alarm rate is a measure of the internal noise of the PA11 and is strongly dependent on the discriminator level. The peak value of the false alarm rate occurs at the voltage where the comparator changes logic states, the threshold voltage. Normalized False Alarm Rate. The normalized false alarm rate is defined as the ratio of the false alarm rate at a particular discriminator setting to the maximum rate to which the system can respond. For example, if this maximum rate is 100 MHz, and the sensitivity is specified for a normalized false alarm rate of lo4, then the sensitivity is measured at the discriminator threshold voltage which gives a false alarm rate of 100 Hz. Sensitiuity. The sensitivity is the pulse amplitude for which the output frequency of the PAD is equal to one half the frequency of the input signal (Le., the level of the signal a t which one half the pulses are detected and one half are not detected). If no normalized false alarm rate is given with a sensitivity, then it is implied to be lo4. The combination of sensitivity and normalized false alarm rate provides a uniform method of comparing PADS that have different internal noise levels and response rates. Pulse Pair Resolution (PPR). We define the PPR as the period of time between the trailing edge of the leading pulse to the leading edge of the trailing pulse (both at the half-maximum) at the point where the frequency of the output signal is 1%less than the frequency of the input signal. This term is defined so as to minimize the effect the pulse width has on the PPR; however, the PPR will still be affected to some extent by the pulse width. Design Goals. The PAD described in this article is intended to be useful with a variety of analytical instruments which utilize photomultipliers and electron multipliers. In order for the PAD to be useful for most applications, the primary design criteria were chosen to be sensitivity and pulse pair resolution. Sensitiuity. It is possible to estimate the approximate sensitivity needed for a particular PMT, once its gain and pulse width are known, if one assumes a triangular shape for the pulse (10). The gain varies widely from lo4 to los, with 5 X lo6 being typical, and the pulse widths vary from 2 to 30 ns, with 10 ns being typical (16,17). The approximate peak amplitude of the pulse is A,, G is the gain of the electron multiplier, Wp is the Full Width of the pulse at Half-Maximum (FWHM) amplitude, e is the charge on an electron in Coulombs, and RL is the anode load resistor (typically 50 a): A, =
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RL (1) WP Substituting the typical values into this equation gives the result
t 2 1980 American Chemical Society
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Figure 1. Block diagram of the PADIcounting system
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Figure 2. PAD schematic. All resistances in ohms: all capacitances in microfarads. R 1 and R2 are discussed in the text
that a typical pulse from a P M T has a peak value of 6 mV. For PMTs with higher gains or narrower pulses, the peak voltage will be higher and for those with lower gains or wider pulses, the peak voltage will be lower. We have observed pulses smaller than 1 mV and as large as 30 mV from various PMTs. This dictates that the PAD should have a wide dynamic range to be useful with all electron multipliers. Therefore, the PAD was designed to have a useful sensitivity of at least 500 fiV and function up to 40 mV. The PAD can be used with larger pulses if their amplitude is reduced into this working range by lowering the voltage applied to the P M T or by attenuating the signal. Pulse Pair Resolution. A major design goal was that the PPR should not degrade the overall system performance, and thus should be less than twice the narrowest expected pulsewidth (base line-to-base line) of the electron multiplier (which would be 8 ns for fast electron multipliers). Input Impedance. It is important that the amplifier have an input impedance which matches the cable impedance. The most
commonly used coaxial cables (for short lengths, RG-58/U or for longer lengths, low loss RG-8/U) have a 5 0 4 impedance. An impedance mismatch causes reflections which can result in a single pulse being counted many times, depending on the degree of mismatch at both the PMT/cable and cable/PAD interfaces, and to a lesser extent on cable losses. Thus, the PAD was designed to have a 5 0 4 input impedance. Output Drive Capability. Many designs use ECL logic in applications requiring high speed, long cable driving capability, high noise immunity, and low radiated noise. The comparator (Advanced Micro Devices type AM685 in Figure 2) (18) of the PAD described here uses ECL logic instead of the more common TTL logic for two important reasons. The first is low radiated noise. The comparator has complementary output stages in which both output transistors are always on. When the comparator changes state, one transistor stage turns on more and the other less, thus there is very little current surge to couple back to the amplifier input. In T T L logic, the output stage is either on or off, and changing states creates a current surge which might be
ANALYTICAL CHEMISTRY, VOL. 52, NO. 8, JULY 1980
coupled to the input. The second reason to use ECL devices is that they can drive a 5 0 4 load. This means that the counter can be located remotely from the PMT/PAD without degrading system performance. Therefore, the PAD can be located as close as possible to the PMT, minimizing signal loss in the PMT/PAD cable, and thus maximizing the signal amplitude to the PAD. PAD Description. The PAD is designed to have sufficient gain to cause either a positive or negative going pulse with an amplitude of 500 FV to trigger the comparator. The schematic for the pulse amplifier/discriminator is given in Figure 2. The input signal is developed across an isolation transformer and then passed on to two cascaded transistor differential amplifier stages. The output of the differential amplifiers is the input to the ECL comparator. Amplifier. The input transformer consists of a ferrite bead with a single turn primary winding and a two-turn center-tapped secondary winding. The input impedance of the transformer has been made to match the coaxial cable (50 Q ) by the addition of a small resistor in series with the signal. The input transformer is used for two reasons. The first is to split the input signal into two balanced signals 180’ out of phase. The second is to isolate the rest of the circuit from any dc voltages which may be superimposed on the signal since the transformer passes only ac signals. The first differential amplifier stage has 2 mA of current through each transistor. This current minimizes the amount of noise added by the transistors (19). This is vital since any noise added in the first stage will be amplified by any succeeding stages. For best circuit performance, the collector currents should be matched to within 10% by selection of transistors. The differential output from the first stage is transmitted to the second stage through coupling capacitors. These capacitors transfer only the change in signal to the base of each transistor in the following stage where further amplification occurs. It is advantageous to use capacitor coupling as it effectively forms a high pass filter, blocking any dc drift from interfering with the operation of the next stage. High speed transistors also have problems with l / f noise, and this problem is also minimized by the use of coupling capacitors. The second differential amplifier stage has 5 mA passing through each transistor. The additional current increases the gain of this stage at a small sacrifice in noise performance (19). The lower noise performance of the second stage is not a problem since the amplified noise from the first stage is larger than the noise added by the second stage. Comparator, The signal from the second stage is capacitively coupled to the comparator. There are several advantages of ac coupling the signal to the comparator: (1) the gain of output transistors of the preceding stage do not have to be closely matched, even though any differences in gain will cause the collector voltages to differ; (2) the discriminator level is more stable since any dc drift in the preceding stage is blocked by the capacitors; (3) the signal is passed through a high pass filter which discriminates against l / f noise and long term drift; and (4)the comparator performs best when the common mode voltage is close to ground (18)which is obtained by referencing the output signal to ground through a resistor network. The discriminator level may be adjusted over a 100-mV range and the center of the range may be changed by varying the ratio of R1 to R2. The sum of R1 and R2 should equal about 200 Q. The discriminator level is available as a BNC output. This output has been filtered to keep noise injected onto it from the voltmeter from affecting the comparator. Shorting the discriminator output, although not recommended, will not harm the circuit or significantly affect circuit operation. The comparator used is essentially a high speed (>300 MHz), high gain (>1600) amplifier (18) designed to transform analog signals into digital signals. The comparator has complementary outputs available and the use of either or both outputs will not affect circuit operation. These outputs are each capable of driving a 5 0 4 load (coaxial cable) for considerable distances. It is required that the outputs be terminated properly and the terminations be as close as possible to the input of the device(s) being driven. Counter Description. The schematic for the counter is given in Figure 3. The counter described here uses a high speed ECL front end to ensure that the counter does not degrade the system
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performance. The signal is then converted to T T L for ease of counting and interfacing to a microcomputer after the ECL front end has reduced the signal rate sufficiently that standard TTL integrated circuits ( I C ) may be used. A microcomputer is used to control and process the data from the counter(s). There are several advantages to designing the counter using ECL logic (20). The signal can be brought to the counter via a coaxial cable since ECL devices are able to drive a 5 0 4 load. The use of a coaxial cable reduces the amount of noise radiated from the signal into the counting environment. The high speed signal does not need to be converted into TTL if one uses ECL devices to count it. Noise generated by TTL integrated circuits can cause serious problems a t the high frequencies which may be encountered in photon counting. The use of ECL devices minimizes this problem since ECL devices generate much less noise than TTL devices. The speed of the counting circuitry in photon counting is quite important because if it is not fast enough, it can limit the performance of the whole system. A burst of pulses may occur at an instantaneous rate which is much greater than the average rate because the signal is distributed randomly in time. Advanced T T L logic is limited to counting signals with a frequency of no more than about 100 MHz. T o use TTL counters, the signal must be transformed to TTL pulses. The standard I.C. (10125) designed to perform this task is limited to a signal rate of about 80 MHz, which would limit the overall system performance. A 225-MHz flip-flop (manufacturer’s “typical” specification (21))was chosen to count the signal since it should not limit the system performance. The output of the first flip-flop is cascaded through two more ECL flip-flops to ensure that the signal will be decreased to a rate such that conventional T T L counters can be used. The maximum signal rate after being passed through three ECL flip-flops will be one eighth of the maximum rate that the first flip-flop can toggle. At this maximum rate, about 30 MHz, the signal is converted to TTL. The ECI,/TTL conversion can now be done using a 10125 I.C. since the maximum signal rate has been reduced to a level where it performs adequately. After the signal has been converted to TTL, it is counted using three dual, 4-bit binary counters (74LS393). The largest number this counter can accumulate without overflow is 2B - 1(=8388607), since the 24th bit is used to indicate an overflow condition. The counter is gated using an external time base generator (not described here) which provides a counter reset pulse and controls the time that the first ECL flip-flop can count. The time base can be programmed to provide a counting period from 1F S to 14 h. However, one can count indefinitely by using the microcomputer to store either individual resul1,s of short time segments or their sum. The data from the ECL flip-flops are converted to TTL after the signal is gated off. This conversion is done using the 10125 I.C. without any loss of high-speed performance since the data are no longer changing. The converted data from the ECL flipflops and the binary counters are transferred to a standard peripheral interface I.C. (8255). The microcomputer then reads the data from this I.C. and afterwards generates the counter reset pulse which starts the process again. The microcomputer is not actively involved during the counting process and so is free to perform other functions while the data are being collected. Prescaling the output to decrease the signal frequency has been advocated (22); however, the ECL counter described here is capable of operating faster than the PAD. This makes prescaling unnecessary and removes the uncertainty in the count associated with prescaling.
RESULTS AND DISCUSSION T o measure the detection sensitivity and the pulse pair resolution of the PAD, one must have a pulse train of stable amplitude and frequency with a pulse shape similar t o that of a PMT. In these experiments, such pulses were generated using either a crystal controlled pulse pair generator of our own design t h a t has a continuously variable separation between the pair of pulses and individually continuously variable pulse width and amplitude (23),or with a IIP 8082A pulse generator, which is capable of generating a pulse frequency
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ANALYTICAL CHEMISTRY, VOL. 52, NO. 8, JULY 1980 RESET COUNT
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Figure 3. Counter schematic. All resistances in ohms. Integrated circuits: (A and B) 74LS136, open collector exclusive-or gates; (C) Intel 8255, Programmable peripheral interface; (D, E, and F) 74LS393, dual 4-bit binary counters; (G) 74S00, Schottky NAND gate; (H) 10124, TTL-to-ECL translator; (I) 10131, dual ECL type D master-slave flip-flop; (J) 10231, dual ECL type D master-slave flip-flop; and (K) 10125 ECL-to-TTL translator
t o slightly above 250 MHz. The frequency of the signal from the H P 8082A was measured either by counting the output for a known period of time or by measuring the pulse separation using a Tektronix 7904 oscilloscope (500-MHz bandwidth). Sensitivity Measurements. T o measure the sensitivity of the PAD, the signal from the pulse pair generator was attenuated (using a calibrated 50-0 attenuator) until the frequency from the comparator had decreased to one half the original frequency. The sensitivity function is quite sharp a t the 50% point. At this point, a change in the signal magnitude from +1 d B to -1 dB can cause about a 50% reduction in the observed pulse rate. T h e results of the PAD sensitivity measurements are given in Figure 4 for 3.5-11s wide pulses (base line-to-base line) or 2 ns FWHM. T h e detection sensitivity of the PAD as a function of the discriminator voltage is reasonably flat until the discriminator level begins to approach the area where amplifier noise is significant. T h e amplifier false alarm rate is also plotted showing that its noise is confined to small values of the discriminator voltage. The ultimate sensitivity of the PAD for a single pulse is e100 pV with a normalized false alarm rate of lo* (=120 Hz) or =250 pV for a normalized false (=1 Hz). Figure 5 gives the same inforalarm rate of mation for the pulse amplifier/discriminator designed by Darland et al. (22) and a commercial amplifier and discriminator. T h e single pulse detection sensitivity for the PAD (100 FV) is significantly better than t h a t of either Darland (600 pV) or ORTEC (900 pV). Darland uses a video amplifier I.C. that has a variable gain. T h e curves shown are for the minimum
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and maximum gain. Changing the gain has negligible effect on the detection sensitivity because it also changes the amplifier noise and hence the false alarm rate. T h e pulses used to test the detection sensitivity and P P R were 2 ns wide (FWHM) so that the width of the pulse would
ANALYTICAL CHEMISTRY, VOL. 5 2 , NO. 8, JlJLY 1980
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Discriminator Setting (arbitrary units1 50 130 90 108
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Figure 5. Detection sensitivity and false alarm rate vs. discriminator threshold for Darland’s design (22) and the ORTEC amplifier (model 454) and discriminator (model 436). The ORTEC (0)and, for Darland, the minimum gain setting (A)and maximum gain setting ( 0 ) . The discriminator setting of the ORTEC corresponds to the number of turns for a potentiometer (full scale = 1000)
not limit the ability of the system to respond. The effect of the width of the pulse on the detection sensitivity of our PAD was determined since many PMTs have pulse widths that are greater than 2 ns. T h e detection sensitivity of the PAD was found t o halve from 2 to 15 ns (FWHM) and then to increase slightly for pulses up t o 50 ns (FWHM). P u l s e Pair R e s o l u t i o n M e a s u r e m e n t s . T h e ability of the PAD to distinguish two separate pulses is a function of the amplitude, separation, and width of the pulses. The P P R is measured using a pair of pulses which are relatively close together (from 0 to 100 ns) and repeated a t relatively long intervals (here, 1000 ns or a t 1 MHz). For these measurements, the trailing pulse of the pulse pair was brought closer to the fixed pulse until the frequency measured decreased by 1% . T h e interval between the trailing and leading edges of these two pulses (measured at the half-maximum) is the pulse pair resolution. Plots of the pulse pair resolution vs. pulse amplitude for the three systems a t a false alarm rate of 100 Hz are given in Figure 6. T h e P P R and dynamic range of the PAD (7 ns a n d 350 g V to >40 mV) is better than t h a t of ORTEC (8.5 ns and 1.4 mV to >40 mV) and significantly better than that of Darland (12 ns and 1.0 mV to 2.0 mV). The variable gain feature of Darland’s design produces a family of curves similar to the one shown with decreasing sensitivity and dynamic range, and improving P P R (ultimately reaching a P P R of 8 ns with a 7-mV pulse and a dynamic range
