ANALYTICAL CHEMISTRY, VOL. 51, NO. 2, FEBRUARY 1979
to be generalized; rather, they serve to demonstrate than even systems which use the same pulse counting electronics can behave quite differently, and that it is extremely important to characterize the behavior of any give pulse counting system to ensure that it does in fact satisfy the measurement requirements.
ACKNOWLEDGMENT We thank S. R. Crouch for many helpful conversations during the preparation of this manuscript.
245
(6) F. T. Arecchi, E. Gatti, and A. Sona, Rev. Sci. Instrum., 37, 942 (1966). (7) F. Robben. Appl. Opt., 10, 776 (1971). (8) J. D. Ingle, Jr., and S. R . Crouch, Anal. Chem., 44, 777 (1972). (9) L. Birenbaum and D. B. Scarl, Appl. Opt., 12, 519 (1973). (IO) D. C. Wenke, Ph.D. Thesis, Michigan State University,East Lansing, Mich.. 1972. (11) M. VanGorkom and R . E. Glick, Int. J . Mass Spectrom. Ion Phys., 4, 203 (1970). (12) M. VanGorkom, D. P. Beggs, and R . E. Glick, I n t . J . Mass Specfrom. Ion Phys., 4, 441 (1970). (13)J. M. Hayes and D. A. Schoeller, Anal. Chem., 49, 306 (1977). (14) J. M. Hayes, D. E. Matthews, and D. A. Schoeller, Anal. Chem., 50, 25 (1978). (15) C. Smit and C. Th. J. Aikemade, Appl. Sci. Res., 108, 309 (1963-64).
LITERATURE CITED (1) J. D. Ingle, Jr., and S. R. Crouch, Anal. Chem., 44, 785 (1972). (2) K . C. Ash and E. H. Piepmeier, Anal. Chem.. 43, 26 (1971). (3) E. J. Dadand, PkD. Thesis, Michigan State University, East Lansing. Mich., 1978. (4) E.J. Dadand, J. E. Hornshuh, C. G. Enke, and G. E. Leroi, Anal. Chem., (5)
following paper in this issue. L. Heroux, Appl. Opt., 7, 2351 (1968).
RECEIVED for review July 14,1978. Accepted October 30,1978. Two of the authors (E.J.D. and G.E.L.) are pleased to acknowledge the support of the National Science Foundation (MPS 75-02525) and the Office of Naval Research (N00013-76-C-0434).
Pulse (Photon) Counting: A High-speed, Direct Current-Coupled Pulse Counter E. J. Darland, J. E. Hornshuh,' C. G. Enke," and G. E. Leroi Department of Chemistry, Michigan State University, East Lansing, Michigan 48824
An amplifier/discriminator/prescaler module which forms a complete, high-performance pulse counter when combined with any standard TTL counter is described. It is constructed with readily-avallable Integrated circuits and costs less that $150, yet has high sensltlvlty (130 p V ) and is capable of very high count rates (>90 MHz with periodic Input pulses; pulse pair resolution = 11 ns). The entire circuit is dc coupled so that there is no base-line shift at any count rate. The discriminator level is voltage programmable and the amount of prescaling can be selected remotely. Extra precautions taken to ensure drift-free operation of the electronics result in a pulse counter with excellent long-term stability.
I t is widely recognized that pulse counting is the best method for measuring the output signal from electron multipliers used as particle (photon, ion, electron, etc.) detectors when the incident particle flux is very low (1-8). Likewise, it is recognized that pulse overlap counting losses inherent in the pulse counting technique limit its accuracy at higher particle fluxes, so that direct current (dc) techniques are best used at these higher fluxes (5-9). In spite of the fact that dc measurement systems which are adequate for mediumand high-flux measurements are readily available, there are several practical reasons for the continued interest in developing pulse counting systems which can operate at ever higher count rates. For example, pulse counting systems may have significant signal-to-noise ratio and stability advantages over dc systems at equivalent fluxes (3-8). Furthermore, pulse counting is an inherently digital technique; pulse counting measurements require fewer data domain conversions (10) Present address: Central Research Department, Experimental Station, E. I. du P o n t de Nemours, Wilmington, Del. 19898. 0003-2700/79/0351-0245$01.00/0
than dc measurements, and are thus less susceptible to the inevitable errors which accompany such conversions ( 8 , I O ) . The development of faster pulse counting systems not only extends these advantages to measurements made a t higher fluxes, but also has important consequences in terms of cost and convenience. The availability of a pulse counting system which is as fast as possible increases the probability that all necessary measurements (or a t least an entire series of measurements) can be made without resorting to dc techniques. Thus, in some applications, the cost of a dc system may be eliminated entirely. Even when dc measurements must be made, a fast pulse counting capability will reduce the sensitivity requirements for the dc system. The pulse counter described here has performance characteristics approaching the practical limits, but it can be built for a small fraction of the cost of present commercial pulse counting systems. The pulse counter has high sensitivity as well as stable and easily adjusted gain and discriminator levels, which are independent of pulse rate. The discriminator level is voltage programmed, and the amount of prescaling can be selected remotely. These features allow the possibility of real-time computer control of the measurement system parameters, which could be automatically optimized to meet changing experimental conditions (11).
PULSE COUNTER DESIGN A block diagram of the pulse counter is shown in Figure 1. The digital counter is a standard TTL design, interfaced to a PDPB/I minicomputer, and the clock is the real-time clock of the computer (12). These units are not described in this paper. For the sake of brevity, the remainder of the circuitry (amplifier, discriminator, prescaler, and line driver) will henceforth be termed the "pulse counter", although it should be realized that some sort of frequency meter (or digital counter and clock) are also necessary to make a complete pulse counter. 0 1979 American Chemical Society
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ANALYTICAL CHEMISTRY, VOL. 51, NO. 2, FEBRUARY 1979
+8
D I G I T A L COUNTER
T V TO
OR
+2
+4
LINE
+I
DRIVER
1
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D I S P L A Y COMPUTER
Figure 1. Block diagram of the pulse counter
Establishment of Performance Goals. The pulse counter described here was intended for use with a wide variety of computer-interfaced spectrometric instrumentation, a t both very high and very low count rates. The primary design goals were high speed, sufficient sensitivity for use with common photomultiplier tubes, and sufficient stability for low count rate experiments lasting as long as several days. The speed of the counter is important since pulse overlap in the counter circuits is usually the limiting factor in determining the maximum count rate at which accurate data can be obtained (9). State-of-the-art counters are capable of counting periodic pulses up to about 100 MHz, i.e., they can resolve pulses separated by as little as 10 ns. However, particles usually arrive a t the electron multiplier with a random time distribution. In order to measure the average pulse rate accurately, the pulse counting electronics must be capable of responding a t an instantaneous pulse rate which is greater than the average pulse rate by a factor which depends on the amount of pulse overlap counting loss that can be tolerated. The average pulse rate for which good accuracy can be obtained is only a few percent of the maximum count rate for the counter. Still faster pulse counting electronics might be expected to help the pulse overlap problem significantly. However, even with relatively fast electron multipliers, the full-width at half-maximum (FWHM) of the pulses observed at the counter input will be on the order of 2-3 ns. For most pulse counting, the best stability and signal-to-noise ratio will be obtained when the discriminator is set significantly below half of the mean pulse amplitude (11). Under these conditions, the mean pulse width may be significantly larger than the FWHM, or on the order of 6-10 ns. Thus, a 100-MHz pulse counter with a 10-ns dead-time approaches the practical limit set by pulse overlap within the electron multiplier and connecting cable, and a design goal of 100 MHz was adopted. The sensitivity of the counter is important for two reasons. Pulse heights from a typical electron multiplier have a random and broad distribution. If the counter is not sensitive enough to count the smallest pulses, then some of the signal is wasted, increasing the measurement time or lowering the signal-tonoise ratio of the measurement (8). Furthermore, although the discriminator setting a t which the best overall system stability is attained will vary from system to system ( I I ) , the optimum setting is likely to be near the “small” end of the pulse height distribution, so that the pulse counter should have sufficient sensitivity to count nearly all of the pulses. Ease of use and restricted space within most photomultiplier
housings dictated that the pulse counter not be mounted inside the housing, but rather in a separate nearby enclosure connected to the electron multiplier or photomultiplier with the shortest practical length of common 50- (or loo-) Q coaxial cable. T o minimize reflections, an anode load resistor of 50 (100) 0 must be used a t the pulse amplifier input. The amplitude of the voltage pulses generated across this resistor by the anode current pulses determines the sensitivity required of the rest of the circuit. We performed pulse height measurements on several multipliers (RCA 1P28 and 8850 PMTs and a variety of “bare” electron multipliers used as mass spectrometer ion transducers). The multipliers which had the lowest gain a t normal operating voltages produced pulses which were as large as 5-10 mV (50-f2 load), but also many pulses which were smaller than 1 mV. Therefore, a design goal of a t least 500 pV sensitivity was adopted. The pulse counter described here uses a fast binary counter between the discriminator and line driver to prescale the discriminator output, as depicted in Figure 1. The true number of counts must then be obtained by multiplying the recorded number of counts by the same factor by which the prescaler divides the pulse rate. Inclusion of prescaling has two very desirable effects. The first is that the average pulse rate at the input to the line driver is reduced by the scaling factor, so that succeeding circuitry does not have to be so fast. The second, and even more important, advantage is that prescaling reduces the probability of pulse overlap by more than it reduces the average pulse rate. This can be illustrated as follows. The probability that n pulses will occur within the resolving time t of a pulse counter is given by the Poisson distribution as
where R is the average pulse rate. If zero or one pulse arrives during t , no pulse overlap occurs, but if two or more arrive during the same interval, pulse overlap counting loss has occurred. The probability of this happening is then 1
Po = 1 -
c P(n,t)
n=O
If, however, the input pulses are prescaled by two (and the prescaler is so fast that no pulse overlap loss occurs in it), then three or more pulses must arrive at the prescaler input within
ANALYTICAL CHEMISTRY, VOL. 51, NO. 2, FEBRUARY 1979
5 MHz
20W. CM,.,
Po=00366
Figure 2. Probability of pulse overlap in the counter ( P o ) for four combinations of prescaling factor and maximum counting rate
the resolving time of the counter to cause pulse overlap in the counter, and
Po = 1 -
c P(n,t)
n=O
In general, for a prescaling factor I , I
Po = 1 -
c P(n,t)
n=O
The advantages of prescaling are illustrated in Figure 2. Note that, although the ratio of the pulse rate at the digital counter input to the maximum frequency of the digital counter is constant in the four examples in Figure 2, Pois reduced by a factor of more than 100 when a divide by eight prescaler is introduced. The only disadvantage of prescaling is that the count resolution of the readout is lowered by the prescaling factor, so that longer integration times may be required to reach the desired count total and/or precision. This problem could be serious a t low count rates except that, at low count rates, less prescaling is required. Therefore, three levels of prescaling are provided (in addition to the “no prescaling” or +l option) so that the count resolution can be maintained as high as possible while keeping the maximum frequency within a range that the line driver and succeeding circuitry can handle. Amplifier/Discriminator Circuit. The preamplifierdiscriminator portion of the pulse counter is very simple; it is based on only two integrated circuits, as shown in Figure 3. A high-speed comparator (Advanced Micro Devices type
AM685) with differential inputs and emitter-coupled logic (ECL) outputs is used as the discriminator. The amplifier is a Texas Instruments ~ A 7 3 3adjustable gain, high-speed differential video amplifier. When the counter is in its quiescent state (both amplifier inputs at essentially zero volts), the amplifier outputs have a common mode voltage, VA,,, which is typically about 2.9 V, and are separated by an offset voltage, VAo,which depends on the gain of the amplifier, but is typically 0.6 V a t the maximum gain of 400. These voltage levels are shown in the amplifier waveform plots of Figure 4. A negative pulse at the input of the amplifier causes the outputs to move closer together, and for sufficiently large pulses, the outputs will cross. A quiescent comparator input common mode voltage, Vc,,, and an input offset voltage, VC,, can be defined which are analogous to VA,, and VAo. The comparator is triggered whenever its inputs cross, i.e., when the input pulse to the comparator is larger than Vco. Capacitive (or ac) coupling between the amplifier and comparator would allow Vc,, and Vco to be adjusted independently of VA,, and VAo. However, ac coupled circuits exhibit base-line shift which changes the effective threshold level at higher count rates (5). The various diode clamp techniques which are commonly used to limit base-line shift do not completely eliminate it ( I 3 ) ,and design of base-line correction circuits suitable for use at 100 MHz is not a trivial problem. Nor is simple direct coupling between the amplifier and comparator a possibility, since an amplifier gain of 1200 would be required if a 5 0 0 - ~ Vsignal were to cause threshold crossing. This pulse counter retains dc coupling, but through two simple resistor networks as shown in Figure 3. Resistors R3 and R4 (and similarly R6 and R7) act as simple voltage dividers, transmitting a large fraction of the amplifier output pulse to the comparator input. However, R5 and R8 allow the voltage a t one end of each divider to be varied independently over a range of +6 V to -6 V, which in turn allows independent control of the quiescent values of the comparator input voltages. With the component values shown, Vc,, can be reduced to approximately 1 V, which results in much improved overload characteristics. The comparator offset voltage, VC,, can be adjusted continuously within a range of
UPPER LEVEL ADJUST
INPUT
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circuit.
All
247
resistances in ohms: all capacitances in microfarads.
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ANALYTICAL CHEMISTRY, VOL. 51, NO. 2, FEBRUARY 1979
source impedance seen by the comparator a t its inputs ( 1 4 ) . Careful construction techniques can help minimize the stray capacitance between comparator output and input. The use of a comparator with ECL outputs is also advantageous, since the voltage swing for an ECL logic transition is only about one fifth of that for a T T L logic transition, and thus there is less signal to couple back to the inputs. The width of the critical range is also found to be smallest when the Comparator is driven by a balanced source with an output impedance of about a thousand ohms or a little larger. Coupling resistors R3 and R6 also serve t o establish the desirable source impedance. The inclusion of R3 and R6 has one major disadvantage. Every pulse from the amplifier must charge up the input capacitance of the comparator through these resistors, and if too large a resistance is used, the resulting RC time constant becomes the limiting factor in the frequency response of the counter. The value of 680 ti' was chosen as a compromise between high frequency response and small critical range (Le., high sensitivity). Different values for these resistors could improve either of these characteristics at the expense of the other. Prescaler/Linedriver Circuit. The prescaling and line driver circuitry of the pulse counter is shown in Figure 5. The signal from the comparator is converted into a TTL level signal (+I) which is used to drive the light emitting diode (LED) that indicates the state of the comparator (it is on when the comparator is in its quiescent state). Since R5 and R8 (Figure 3) can be adjusted to make Vc, negative (so that the comparator is permanently triggered), the STATE LED is particularly useful when adjusting the threshold level of the pulse counter. The output of the comparator also goes to a binary counter composed of three fast (140 MHz) ECL flip-flops which divide the pulse stream from the comparator by 2, 4, and 8. The output from each stage of the counter is also converted into a T T L level signal. A 74153 multiplexer passes one of these signals (or the +1 signal) to the 75109 differential current
'-oo5j - 010 Figure 4. Relationship between amplifier input and output voltages Note different scales for input a n d output waveforms
approximately 3~300mV. A typical value of 50 mV results in a tenfold gain in sensitivity compared with the directly coupled amplifier and comparator. Both V,,, and Vcocan be varied independently of the amplifier gain, which can be set so as to obtain the best compromise between bandwidth and temperature stability. Furthermore, there IS n o base-line shift a t an3 count rate. T h e comparator is essentially a high-gain, high-speed amplifier, and as such is particularly prone to oscillation. Oscillations will occur when the differential input voltage remains in some critical range near zero for longer than the propagation delay of the comparator (14). Therefore, the input signal must traverse the critical range in a time period which is less than the propagation delay of the comparator if oscillation is to be avoided. Fortunately, the rise time of the amplified anode pulses is much shorter than the comparator propagation delay time, so that hysteresis (which can only limit sensitivity and maximum speed) is unnecessary. T h e tendency to oscillate does limit the overall sensitivity of the counter, however, since the critical range limits the minimum useful value of of Vco. The width of the critical range (which may be as small as a few millivolts or as large as several hundred millivolts) depends on the (fixed) gain of the comparator, the extent to which the comparator's output is coupled back to its input via stray capacitance, and the
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PrescaIerAinedriver circuit. Values of R9 and C11 are not critical (seetext). R l O - R l 7 are 5042 pulldown resistors, R18-R22 are 3-kQ, and R23 and R24 are each one-half the nominal impedance of the balanced line used to connect t h e line driver and reciever. Dl-D6 are 1N4001 or equivalent Figure 5 .
ANALYTICAL CHEMISTRY, VOL. 51, NO. 2, FEBRUARY 1979
mode line driver, which transmits the pulse train to the digital counter, The line receiver circuit (used a t the input of the digital counter) is shown in an inset in Figure 5 . The signal selected by the multiplexer also triggers a monostable which flashes the PULSE LED, The output pulse width of the monostable is adjusted with R9 and C11 so that the LED emits a visible flash (a few milliseconds long), which is used to indicate whether pulses are being detected by the counter. T h e amount of prescaling used (i.e., which signal is transmitted by the multiplexer) can be determined either manually or under computer control. When the AUTO line is HI, the manual prescale select switches, SW1 and SW2, control the multiplexer. When the AUTO line is LO these switches are disabled and the multiplexer is controlled instead from the computer via the remote prescale select lines. Two LEDs are used to indicate the status of the multiplexer control lines. It should be noted that the output pulses from the discriminator are so narrow (- 11 ns) that the “+1” output of the prescaler does not trigger the line driver reliably, and single-count resolution should normally be obtained by counting both positive- and negative-going edges of the + 2 output ( 1 5 ) . The entire circuit of the photon counter is designed so that only two main supply voltages (+6 V and -6 V) are needed. T h e auxiliary voltages needed for the T T L and ECL integrated circuits are derived on the circuit board using the simple diode circuits also shown in an inset in Figure 5. Construction. When working with high frequency, low level signals, the construction methods used are as important as the rest of the circuit design, and some of the details of construction found to be necessary with the pulse counter are summarized here (16). The entire circuit (except for the power supply) is built on a single, 4 inch X 9 inch, double-sided printed circuit board, on which as complete a ground plane as possible is maintained. Since the amplifier section of the circuit is very sensitive to high frequency signals, extra precautions are necessary to shield the amplifier and comparator from both internal and external noise sources. For instance, fast logic edges (such as normal T T L transitions) are very rich in frequencies above 30 MHz, which can be radiated very easily. I t was found necessary to place a metal shield across the circuit board between the amplifier-comparator section and the ECL and TTL logic of the prescaler section to prevent noise radiated in the latter section from being picked up a t the amplifier input. Better results were obtained when the input and output connectors (as well as the PULSE and STATE LEDs) were soldered directly to the circuit board instead of being connected with short wires, since such wires radiate and pick up unwanted noise more easily than does foil on a circuit board with a good ground plane. The main circuit board is shielded by internal partitions from the ac voltages present in both the power supply and temperature control circuits. The dc power to the circuit board is passed through the partitions via small feed-through capcitors which are used to bypass high frequency noise. Ferrite bead and capacitor filters are used to filter any remaining high frequency noise from the power inputs to both the amplifier and the comparator (see Figure 3).
Not only must the pulse counter be mounted in an enclosure which shields it from external radiated noise, but extreme care must be taken to ensure that no noise “sneaks in” on the 115-V ac power line. The pulse counter includes, a t the power entrance to the enclosure, both a radio frequency interference (RFI) filter and transient suppressors which successfully eliminate such noise. Preliminary experiments indicated that the thermal characteristics of the circuit could be important. The pulse
249
Figure 6. The completed pulse counter
counter was therefore built in an enclosure which was large enough to include a small circulating fan and proportional temperature control circuitry so that the thermal behavior of the counter could be easily studied and regulated. The completed pulse counter (shown in Figure 6) has overall dimensions of 11X 3.5 X 7.5 inches. The total cost for all parts and materials in the pulse counter unit was less than $130. PULSE COUNTER PERFORMANCE Sensitivity. It is necessary to calibrate the sensitivity of a pulse counter using pulses with about the same width as the actual anode pulses from the multiplier to be used. The FWHM of the anode pulses from the 1P28 are about 5.5-6 ns; the pulses from the 8850 and the faster “bare” multipliers were similar a t moderate voltages. The narrowest pulses obtainable from a Tektronix PG-501 pulse generator had a FWHM of just 5.5 ns, and an amplitude which was variable over a range of about 0-5 V. These pulses were connected to the counter via a calibrated 66-db attenuator, which reduced their amplitude to a range of 0-2.5 mV. The input pulse amplitude necessary to cause reproducible triggering of the pulse counter (with a 50-fl load resistor) was measured as a function of discriminator setting. The lower limit of sensitivity, determined by the onset of comparator oscillation, is about 130 pV. In practice, threshold settings lower than about 250 p V are not used, so that numerous small noise pulses which are picked up by the cable connecting the counter to the multiplier will not be counted. Measurement of V,, a t the same threshold settings indicated that the effective gain of the amplifier for such narrow pulses is about 250, although the low frequency gain is nearly 400. Speed. Pulse counter speeds are specified in several ways. One method is to specify the maximum frequency of a periodic waveform to which the pulse counter will respond. The waveform is often not specified, but if it is a series of very narrow pulses, then this method is equivalent to specifying the dead time or pulse pair resolution of the counter. The dead time can also be evaluated by measuring the width of the comparator output pulse for very narrow input pulses (assuming that the amplifier and/or comparator limit the speed of the counter). To specify that a counter is a “100-MHz counter” or that it has a “10-ns deadtime or pulse pair resolution” does not, of course, predict how fast a complete pulse counting system which uses that counter will be, since the behavior of such a system depends critically on numerous other factors (6, 8, 9, 1 1 , 17). However, such specifications can be a useful basis of comparison between different counters of comparable sensitivity, cost, and stability, and such measurements are therefore worth making. The sensitivity of this counter for 5.5-ns pulses showed no
250
ANALYTICAL CHEMISTRY, VOL. 51, N O . 2, FEBRUARY 1979
measurable change up to a pulse repetition rate of 57 MHz, which is the limit of the pulse generator used. The output pulse of the comparator when the amplifier was driven by such pulses was about 11 ns wide. The maximum frequency a t which the counter would trigger reliably with a sinewave input was approximately 93 MHz, which is consistent with a pulse pair resolution of about 11 ns. The speed of several pulse counting systems incorporating this counter is examined experimentally in a companion paper (11). Stability. Some of the results of preliminary experiments using this pulse counter circuit indicated that while the sensitivity was apparently sufficient to count all pulses when used in a photon counting system, the importance of good stability in the pulse counter electronics should not be neglected (11).An examination of the sources of instability in the two photon counting systems described in reference 11 revealed two major contributions. One was a low frequency variation in the light source output of some 170.By far the worst type of instability exhibited by the electronics of the pulse counting measurement system itself was the variation in effective threshold with temperature. Several factors contribute to thermal drift of the effective threshold, but changes in the gain of the amplifier and in VAo(see Figure 4) were found to be most important. Direct measurement of the temperature coefficients of amplifier gain and offset voltage is extremely difficult because the changes are small and because so much low amplitude noise is introduced via the test probes that measurements anywhere near the (nonperturbed) critical range are impossible. A more sensitive (and perhaps more meaningful) method of measuring the change in effective threshold is simply to monitor the count rate a t a "constant" light intensity as the temperature of the counter is varied. Such measurements were performed for several amplifier gain settings. The dependence of count rate on temperature was worst a t low amplifier gain (G = 60), exhibiting a 30% change from 30 to 50 "C. Much better results were obtained a t full amplifier gain (the normal operating conditions), where any count rate changes between 32 and 40 "C were so small as to be totally obscured by the
instability in the light source. The temperature controller used with the pulse counter maintains the temperature to within *0.5 "C indefinitely after an hour's equilibration time. The present data are insufficient to show whether such control is actually necessary for good stability. However, with this temperature control the instability of the counter is almost certainly negligible compared to light source instabilities, or to changes in photomultiplier cathode sensitivity with temperature (18).
ACKNOWLEDGMENT The authors are grateful to B. K. Hahn for suggesting the use of the pA733 and AM685 integrated circuits. LITERATURE CITED F. T. Arecchi. E. Gatti, and A. Sona, Rev. Sci. Instrum.. 37, 942 (1966). A. T. Young, Appl. Opt., 8, 2431 (1969). F. Robben, Appl. Opt., 10, 776 (1971). M. K. Muphy, S.A. Clybum, arid C. Veillon, Anal. Chem., 45, 1468 (1973). M. L. Franklin, G. Hotlick, and H. V. Malmstadt, Anal. Chem., 41, 2 (1969). K. C. Ash and E. H. Piepmeier, Anal. Chem., 43, 26 (1971). H. W. Wemer, H. A. M. m e f t e , and J. V. D. Berg, Int. J. Mss Spectnm. Ion Phys., 8, 459 (1972). J. D. Ingle, Jr., and S. R. Crouch, Anal. Chem., 44, 785 (1972). C. Smit and C. Th. J. Alkemade, Appl. Sci. Res., 108, 309 (1963-4). C. G. Enke, Anal. Chern., 43 ( l ) , 69A (1971). E. J. Darland, G. E. Leroi, and C. G. Enke, Anal. Chem., preceding paper in this Issue. E. J. Darlarid, Ph.D. Thesis, Michigan State University, East Lansing, Mich.,
1978. L. B. Robinson, Rev. Sci. Instrum., 3 2 , 1057 (1981). "Designlng with High-speed Comparators", AMD Application Note, Dec. 1975, Advanced Micro Devices, Sunnyvale, Calif., 94086. K. G. Harf, ComputerDes., 17, 130 (June, 1978). Additional, more detailed, construction information can be obtained by writing the authors. J. M. Hayes, D. E. Matthews, and D. A. Schoeller, Anal. Chern., 50, 25
(1978). Photomultiplier Manual, Technical Series PT-61, RCA Corp., Harrison, N.J.
RECEIVEDfor review July 14,1978. Accepted October 30,1978. Two of the authors (E.J.D. and G.E.L.) are pleased to acknowledge the support of the National Science Foundation (MPS 75-02525) and the Office of Naval Research (N00014-76-C-0434).
Micromolar Voltammetric Analysis by Ring Electrode Shielding at a Rotating Ring-Disk Electrode Stanley Bruckenstein" and P. R. Gifford Chemistry Department, State University of New York at Buffalo, Buffalo, New York
Ring electrode Shielding at a rotating rlng-disk electrode (RRDE) provides a sensitive solid electrode technique for the study of mlcromolar solutions of electroactive materials. At micromolar levels, shielded ring electrode currents are free of nonconvectlve dtffusion current complications which obscure faradaic rotating disk electrode currents. Ag( I), Bi( III), Cu( II), and Fe(II1) and mixtures of Ag(1) and Cu(I1) can be determined in the concentration range 0.1 to 10 X lo-' M.
In the determination of micromolar levels of electroactive species at solid electrodes, the convective diffusion-controlled current for the electrode reaction of analytical interest can
14214
be obscured by charging currents and/or surface processes occurring a t the electrode. This problem can be minimized by use of the rotating ring-disk electrode (RRDE) in the ring shielding mode (1-3). The RRDE has been shown to be of great utility in the study of metal ions in solution since the ring electrode can monitor electroactive species generated or consumed a t the disk ( 4 ) . Moreover, if the ring electrode is held at a fixed potential, complications due to charging and surface processes at the disk do not affect the ring current, thus making the ring electrode shielding technique a sensitive measure of the current due to the disk electrode process. In this study, ring shielding at a RRDE was applied to the determination of Cu(II), Bi(III), Ag(I), and Fe(II1). Simul-
0003-2700/79/0351-0250$01.00/00 1979 American Chemical Society
